Control messaging of common phase error (cpe) compensation performance level for frequency division multiplex (fdm) symbols in wireless communication systems

ABSTRACT

A wireless base station or user equipment (BS or UE) includes a receiver configured to receive a frequency division multiplexed (FDM) symbol transmitted by another wireless BS or UE, and a processor. The processor is configured to process the received FDM symbol to obtain its equalized FDM data subcarriers, generate a CPE estimate using the equalized FDM data subcarriers, and send to the other wireless BS or UE control messages that indicate a CPE compensation performance level using the CPE estimate. The other wireless BS or UE is enabled by the control messages to adapt a density in time and/or frequency of embedded pilot symbols within FDM symbols subsequently transmitted to the wireless BS or UE.

This application is a continuation of U.S. patent application Ser. No.15/940,103, filed Mar. 29, 2018, which claims priority based on U.S.Provisional Application Ser. No. 62/480,806, filed Apr. 3, 2017, each ofwhich is hereby incorporated by reference in its entirety.

TECHNICAL FIELD

The disclosed embodiments relate to mobile telecommunications.

BACKGROUND

The New Radio (NR) definition in 3GPP (3^(rd) Generation PartnershipProject) for mobile systems will encompass a variety of deploymentscenarios envisioned for 5G (fifth generation) mobile communicationsystems. MIMO (multiple input, multiple output) communication systemscan be used for 5G TDD (time division duplex) air interfaces. Flexible(scalable) frame structures are being considered for block symboltransmissions within the new 5G cellular communication standardincluding various frame structure parameters such as FFT (fast Fouriertransform) size, sample rate, and subframe length.

FIG. 1A (Prior Art) is a diagram of an example embodiment including abase station 102 with M antennas 104 that provides a wireless MIMOcommunication system 100. The MIMO base station 102 communicates throughits M antennas 104 with K different user equipment (UE) devices 106,such as mobile handsets, having one or more antennas 108. Spectralefficiency is improved by using massive MIMO communication systemshaving base stations with relatively large numbers of antennas. Exampleembodiments for massive MIMO communication systems are described withinU.S. Published Patent Application 2015/0326286, entitled “MASSIVE MIMOARCHITECTURE,” U.S. Published Patent Application 2015/0326383, entitled“SYNCHRONIZATION OF LARGE ANTENNA COUNT SYSTEMS,” and U.S. PublishedPatent Application 2015/0326291, entitled “SIGNALING AND FRAME STRUCTUREFOR MASSIVE MIMO CELLULAR TELECOMMUNICATION SYSTEMS,” each of which ishereby incorporated by reference in its entirety.

FIG. 1B (Prior Art) provides a diagram of an example time-domainsubframe structure for the LTE (long term evolution) wireless cellularcommunication standard including a common subframe 102 and OFDM(orthogonal frequency division multiplex) symbols 108. For example, withrespect to the 20 MHz (mega-Hertz) bandwidth LTE mode with a normalcyclic prefix 114 and a sampling rate of 30.72 MSps (mega samples persecond), one example subframe structure can be parameterized as follows:

-   -   OFDM symbol 108 length: 2048 samples;    -   common cyclic prefix (CP) 114 length: 144 samples;    -   special cyclic prefix (CP) 112 length: 160 samples    -   multiple OFDM symbols 108 (with CP); and    -   subframe 102 length: 1 ms (millisecond) including 14 OFDM        symbols (with CP).

One objective for the 5G air interfaces is to operate from below 1 GHzto 100 GHz carrier frequencies over a large variety of deploymentscenarios in a single technical framework, for example, using OFDM(orthogonal frequency division multiplexing) modulation. For thisobjective, phase noise (PN) becomes a major impairment at carrierfrequencies above 6 GHz. Phase noise introduces two kinds of impairmenton OFDM-based systems: (1) common phase error (CPE) and (2)inter-carrier interference (ICI). CPE is a common phase rotation acrossall of the subcarriers for an OFDM transmission, and CPE manifests as acommon rotation of the demodulated constellation. The phase noise ateach subcarrier frequency also introduces ICI to the neighboringsubcarriers, and this spectral leakage degrades the orthogonality of theOFDM waveform. This degradation is manifested as a “fuzziness” in eachdemodulated constellation point, and the level of ICI can be measured bythe degradation of the EVM (Error Vector Magnitude) of the communicationlink. Phase noise typically increases with the carrier frequency, forexample, one general assumption is that PSD (power spectrum density)associated with phase noise increases by about 20 dB per decade offrequency.

CPE can be estimated in a straightforward manner with a least squaresestimator according to the equation shown below.

${{\hat{J}}_{0}(m)} = {\frac{\sum\limits_{k \in S_{p}}{{R_{k}(m)}{X_{k}^{*}(m)}{H_{k}^{*}(m)}}}{\sum\limits_{k \in S_{p}}{{{X_{k}(m)}{H_{k}(m)}}}^{2}}.}$

For this equation, R_(k) is the received subcarrier values; X_(k), wherekϵS_(p), is the transmitted pilot symbol that is known at the receiver;H_(k) is the channel estimate; and S_(p) is the subset of thesubcarriers occupied by the pilot. The CPE for each OFDM symbol withinan OFDM transmission is the DC component of the DFT (discrete Fouriertransform) of the baseband PN (Phase Noise) samples over that symbolduration.

As CPE is constant for all subcarriers within an OFDM symbol and can beestimated, CPE compensation can be performed with the introduction ofPhase Noise Reference Signals (PNRS), also called Phase TrackingReference Signals (PTRS), or other pilots within the OFDM transmissions.The addition of the PNRS/PTRS, therefore, allows for CPE compensationbut only at the expense of additional pilot signal overhead within theOFDM symbols. This CPE estimation based on a static pilot pattern,therefore, has the drawback of high overhead due to required pilotsignaling for the purely pilot aided PN compensation. Moreover,different devices and deployment scenarios have different levels ofrequirement for the PN (phase noise) mitigation. For example, UEs (userequipment) and base stations have significantly different phase noisePSD requirements, and UEs can be categorized into different groups withrespect to PN performance based on their frequency band of operation andwireless system application, such as eMBB (enhanced Mobile BroadBand),URLLC (ultra-reliable low latency communications), mMTC (massive machinetype communications), and/or other use cases.

It is noted that the terminology Phase Noise Reference Signal (PNRS) isused herein interchangeably with Phase Tracking Reference Signal (PTRS)to refer to the same signal. In addition to OFDM waveforms, PNRS/PTRScan also be inserted in SC (Single Carrier) waveforms in astraightforward manner to estimate and compensate the complete PN (PhaseNoise) over that SC waveform. Examples of such single carrier waveformsinclude Single Carrier Frequency Division Multiple Access (SC-FDMA), DFTspread OFDM (DFT-s-OFDM), Null Cyclic Prefix Single Carrier (NCP-SC),etc.

PTRS (phase tracking-reference signal) ports and related signals can beused by base stations(s) to allow the UEs to derive a scalar estimationof the common phase error (CPE) due to the phase noise process which isassumed to be constant over all of the subcarriers of a given symbol ofthe allocated UE bandwidth. This estimate becomes more accurate withincreasing the number of REs (resource elements) allocated to PTRSwithin the scheduled bandwidth of the given UE. In addition, whenmultiple antenna ports are used to transmit from the base station(s) tothe UE, there can be one-to-one mapping or many-to-one mapping from theDMRS (demodulation reference signal) ports to the PTRS ports. The DMRSports are used by the base station(s) to provide signals that facilitatedemodulation operations within the UEs. While the use of PTRS ports andrelated communications can help improve CPE compensation, they can alsolead to inefficiencies with respect to the use of available bandwidthand difficulties arise in the selection and allocation of the PTRS portsby the base station(s).

BRIEF SUMMARY

In one aspect, the invention provides a method that includes receiving,by a first wireless base station or user equipment (BS or UE), afrequency division multiplexed (FDM) symbol transmitted by a secondwireless BS or UE, processing, by the first BS or UE, the received FDMsymbol to obtain its equalized FDM data subcarriers, generating, by thefirst BS or UE, a CPE estimate using the equalized FDM data subcarriers,and sending, by the first wireless BS or UE to the second wireless BS orUE, control messages that indicate a CPE compensation performance levelusing the CPE estimate. The second wireless BS or UE, in response to thecontrol messages, is enabled to adapt a density in time and/or frequencyof embedded pilot symbols within FDM symbols subsequently transmitted tothe first wireless BS or UE.

In another aspect, the invention provides a wireless base station oruser equipment (BS or UE) includes a receiver, configured to receive afrequency division multiplexed (FDM) symbol transmitted by anotherwireless BS or UE, and a processor. The processor is configured toprocess the received FDM symbol to obtain its equalized FDM datasubcarriers, generate a CPE estimate using the equalized FDM datasubcarriers, and send to the other wireless BS or UE control messagesthat indicate a CPE compensation performance level using the CPEestimate. The other wireless BS or UE is enabled by the control messagesto adapt a density in time and/or frequency of embedded pilot symbolswithin FDM symbols subsequently transmitted to the wireless BS or UE.

In yet another aspect, the invention provides a non-transitorycomputer-readable medium having instructions stored thereon that arecapable of causing or configuring a wireless base station or userequipment (BS or UE) to perform operations that include receiving, bythe wireless BS or UE, a frequency division multiplexed (FDM) symboltransmitted by another wireless BS or UE, processing, by the BS or UE,the received FDM symbol to obtain its equalized FDM data subcarriers,generating, by the BS or UE, a CPE estimate using the equalized FDM datasubcarriers, and sending, by the wireless BS or UE to the other wirelessBS or UE, control messages that indicate a CPE compensation performancelevel using the CPE estimate. The other wireless BS or UE, in responseto the control messages, is enabled to adapt a density in time and/orfrequency of embedded pilot symbols within FDM symbols subsequentlytransmitted to the wireless BS or UE.

DESCRIPTION OF THE DRAWINGS

It is noted that the appended drawings illustrate only exemplaryembodiments and are, therefore, not to be considered limiting of thescope of the present inventions, for the inventions may admit to otherequally effective embodiments.

FIG. 1A (Prior Art) is a diagram of an example embodiment including abase station with M antennas that provides a wireless MIMO communicationsystem.

FIG. 1B (Prior Art) provides a diagram of an example time-domainsubframe structure for the LTE (long term evolution) wireless cellularcommunication standard including a common subframe and OFDM (orthogonalfrequency division multiplex) symbols.

FIGS. 2-4 provide example embodiments for different implementationswhere clock signals and related circuitry are used as LOs for RFchannels within downlink (DL) and/or uplink (UL) circuitry for basestations or UEs.

FIG. 5 provides an example diagram of two frames transmitted from a MIMObase station that include two orthogonal PTRS ports with the PDSCH.

FIGS. 6A-C provide an example flow diagram for performing the PNcorrelation at the UEs and for the feedback of these correlations fromthe UE to the base station.

FIG. 7 is a block diagram of an example embodiment for a mm (millimeter)wave communication system that can use the disclosed techniques.

FIG. 8 is a block diagram of an example embodiment for circuitry thatcan be used to provide a baseband receiver, a baseband transmitter,multi-FPGA processing circuitry, and a real-time processor with respectto the embodiment of FIG. 7.

FIG. 9 is a diagram of an example embodiment for electronic componentsthat can be used to implement a base station and/or user equipment (UE)for the disclosed embodiments.

FIG. 10 is a block diagram of an example embodiment for symbolprocessing for OFDM transmissions including blind CPE estimation for CPEcompensation.

FIG. 11 is a block diagram of an example embodiment where the CPEestimator includes multiple estimation algorithms including a pilot onlyCPE estimation algorithm, a blind only CPE estimation algorithm, and apilot aided blind CPE estimation algorithm.

FIG. 12 is a process flow diagram of an example embodiment for a blindonly CPE estimation algorithm for the CPE estimator.

FIG. 13 is a process flow diagram of an example embodiment for a pilotaided blind CPE estimation algorithm for the CPE estimator.

FIG. 14 is a diagram of an example embodiment for decision thresholdsfor the different 16-QAM threshold regions described with respect toFIG. 12.

FIG. 15 is a diagram of an example embodiment employing the 8 decisionthresholds for the different 16-QAM threshold regions of FIG. 14 andemploying the embodiment of FIG. 12 method to extract subsets of datasubcarriers of an OFDM symbol that fall within the 8 regions, i.e., thefour real part regions and the four imaginary part regions.

DETAILED DESCRIPTION Terms

The following is a glossary of terms used in the present application:

A spatial stream is a sequence of symbols transmitted from an antennaport. The term spatial stream is used in the present disclosure in thecontext of spatial multiplexing. Spatial multiplexing is a transmissiontechnique used in a MIMO (multiple input multiple output) wirelesscommunication system in which multiple spatial streams are transmittedfrom the multiple transmit antenna ports of the MIMO system.

An antenna port is defined such that the physical channel over which asymbol on the antenna port is conveyed can be inferred from the physicalchannel over which another symbol on the same antenna port is conveyed.For example, an antenna port may convey a DMRS over a channel, and theantenna port may convey on the channel a corresponding spatial streamwhose symbols are demodulated using the channel estimate obtained fromthe received DMRS. For another example, an antenna port may convey aPTRS over a channel, and the antenna port may convey on the channel acorresponding spatial stream whose symbols having phase noise trackedand compensated using the PTRS.

A symbol is a complex-valued signal transmitted over aspace-time-frequency resource. Examples of OFDM symbols are DMRS, SRS(Sounding Reference Signal), data channel symbols and control channelsymbols.

A physical channel is an uplink or downlink physical channel. A physicalchannel corresponds to a set of resource elements carrying informationoriginating from the higher layers of a communication protocol stack.Examples of channels are PDSCH (physical downlink shared channel), PUSCH(physical uplink shared channel), PDCCH (physical downlink controlchannel), PUCCH (physical uplink control channel), and PBCH (physicalbroadcast channel).

A resource element is an element in an OFDM resource grid for an antennaport and subcarrier spacing configuration.

CPE (common phase error) is a common phase rotation across all of thesubcarriers for an OFDM transmission.

Phase noise is partial random phase variation over time in a signal.Typically, phase noise is introduced by non-ideal sinusoidal signalsgenerated by oscillators.

A DMRS is a demodulation reference signal which is used for channelestimation purposes in order to allow demodulation of one or morephysical channels, e.g., PDSCH, PUSCH PDCCH, PUCCH, PBCH.

A DMRS port is an antenna port over which a DMRS is transmitted and canbe used to receive the data symbols transmitted on the same antennaport.

A PTRS is a phase tracking reference signal which is used to track andcompensate for the impact of phase noise. A PTRS is also referred toherein as a PNRS (phase noise reference signal). The describedembodiments advantageously facilitate a reduction in the number of DMRSports (and their corresponding spatial streams) upon which PTRS need betransmitted when CPE correlation between one or more spatial streams isdetected. This is particularly advantageous because PTRS generally needto be transmitted densely in time. Generally speaking, relative to DMRS,for example, PTRS need to be transmitted relatively densely in timebecause phase noise tends to vary more frequently over time than thechannel estimate obtained from the DMRS. Thus, for example, it may besufficient for one DMRS to be sent per slot/TTI; whereas, typicallymultiple PTRS need to be sent per slot/TTI, often within each symbolthereof.

A PTRS port is an antenna port over which a PTRS is transmitted.

A CSI-RS is a channel state information reference signal which is usedto calculate the channel state information.

A CSI-RS port is an antenna port over which a CSI-RS is transmitted.

The disclosed embodiments provide techniques for UEs (user equipment) tomeasure CPE (common phase error) correlations among different receive(or transmit) spatial streams and then to provide feedback to basestation(s) (e.g., gNB) with respect to these cross correlations. For oneexample embodiment, the feedback includes the results of the crosscorrelations such as a correlation matrix or a condensed/transformedversion of it. For one other example embodiment, the feedback includes aselection by the UE of recommended PTRS ports to be associated with itsDMRS ports, and this recommended port list is transmitted back to thebase station(s). The base station(s) (e.g., gNB) then use this crosscorrelation feedback to select and configure the PTRS ports used for oneor more UEs. Other variations can also be implemented while still takingadvantage of the techniques described herein. The described techniquesfor selecting PTRS ports may improve the operation of wirelesstelecommunication systems by enabling them to improve CPE compensationand to more efficiently use available bandwidth.

Example Communication Environments for the Disclosed Embodiments

With respect to the descriptions provided herein, the followingabbreviations are used:

-   -   PTRS: Phase Tracking Reference Signal    -   DMRS: Demodulation Reference Signal    -   TRP: Transmission Reception Point    -   gNB: g NodeB (base station)    -   PSD: Power Spectral Density    -   PN: Phase Noise    -   TTI: Transmission Time Interval, the minimum scheduling interval        for a UE

Example Architectures of Base Station (BS) and User Equipment (UE)

The following are example architectures that can be used for basestation (BS) and user equipment (UE) implementations within a wirelesscommunication system. As these are example embodiments, it is understoodthat additional and/or different architectures could be used.

At the TRP the following are example architectures:

-   -   One antenna panel only        -   One LO (local oscillator) shared between all antennas of the            panel        -   Multiple LOs shared between the antennas of the panel            -   With common clock    -   Multiple antenna panels        -   One LO shared between all antenna panels        -   One LO per antenna panel            -   With common clock shared between all LOs            -   Separate clock for each/some of the LOs        -   Multiple LOs per panel            -   Common clock per panel            -   One clock for all the LOs in that antenna panel

With Multiple TRP joint transmission of some category (Non CoherentJoint Transmission (NCJT), Dynamic Point Selection (DPS), CoordinatedMulti-Point (CoMP), etc.), there are multiple TRPs and/or multiple gNBsthat can communicate to an UE in a coordinated manner. In such cases,the example architecture can have:

-   -   Separate LO(s) per TRP, different clock(s) per TRP    -   Each of the sub options per TRP as given in previous list

At the UE the following are example architectures:

-   -   One antenna panel only        -   One LO (local oscillator) shared between all antennas of the            panel    -   Multiple antenna panels        -   One LO shared between all antenna panels        -   One LO per antenna panel            -   With common clock shared between all LOs            -   Separate clock for each/some of the LOs

In addition to antennas and panels, there are digital transceiverchains, which are mapped to the antennas/panels using an antenna mappingmatrix. Additional and/or different circuitry and components can also beincluded while still taking advantage of the techniques disclosedherein.

FIGS. 2-4 provide example embodiments for different implementationswhere clock signals and related circuitry are used as LOs for RFchannels within downlink (DL) and/or uplink (UL) circuitry for basestations or UEs. It is noted that the LO signals can be used, forexample, to mix RF signals up to higher frequencies and/or to mix RFsignals down to lower frequencies within the DL and/or UL circuitry forbase stations or UEs.

FIG. 2 is a block diagram of an example embodiment where the same clock,which can be generated by oscillator (OSC) 206, and the same PLL (phaselock loop) 204, which can be a divide-by-N based PLL (xN), are used as asingle LO to generate LO signals that are provided to multiple differentRF channels (RF₁ . . . RF_(X)) 202.

FIG. 3 is a block diagram of an example embodiment where the same clock,which can be generated by oscillator (OSC) 206, is used as an input tomultiple different PLLs (PLL₁ . . . PLL_(Z)) 204, which can each be adivide-by-N based PLL (xN). These different PLLs 204 in combination withthe clock are used as different LOs to generate multiple LO signals thatare provided to multiple sets of different RF channels (RF₁ . . .RF_(X), RF₁ . . . RF_(Y)) 202.

FIG. 4 is a block diagram of an example embodiment where multipleclocks, which can be generated by multiple oscillators (OSC₁ . . .OSC_(Z)) 206, are used as inputs to multiple different PLLs (PLL₁ . . .PLL_(Z)) 204, which can each be a divide-by-N based PLL (xN). Thesedifferent PLLs 204 in combination with the different clocks are used asdifferent LOs to generate multiple LO signals that are provided tomultiple sets of different RF channels (RF₁ . . . RF_(X), RF₁ . . .RF_(Y)) 202.

As explained in further detail below, depending on the architectureimplemented for the base stations and/or UEs, the Phase Noise (PN)characteristics are different. Further, the downlink (DL) parts and theuplink (UL) parts of the implementations can also have different PNcharacteristics and should therefore be considered separately.

Example of Different Varieties of MIMO Transmissions

A base station (BS) for a MIMO communication system will set up multiplespatial streams. The following are examples of communication systemswith such multiple spatial streams:

-   -   SU-MIMO (Single User-MIMO)        -   Single TRP        -   Multi TRP    -   MU-MIMO (Multi-User MIMO)        -   Single TRP        -   Multi TRP            For each of these spatial stream examples, the UE            communicating with the base station can be implemented as a            SISO (single input, single output) device or as a MIMO            device.

Also for MIMO systems, there is a mapping between the spatial streamsand the transceiver architectures. To simplify the discussions below,the following assumptions are applied, although it is understood thatthe techniques described herein can be applied to other combinations instraight forward manner.

-   -   SS stands for Spatial Stream; TRX stands for Transceiver chain    -   Mapping of SS to Antennas        -   One SS is mapped within one antenna panel (e.g., the SS is            not shared between the antennas on two or more panels)        -   OR One SS is mapped to multiple antenna panels        -   More than one SS can be mapped to one antenna panel    -   One SS transmission involves only one LO        -   Within a TRP, the SS are mapped to TRX such that each SS has            only one LO    -   Multiple SS can share an LO

While the current NR definition supports up to 12 SS, there arediscussions to extend it to 16 SS (e.g., by supporting 16 orthogonalDMRS ports). The techniques described herein can be scaled up to 16 SSbut are not limited to 16 SS. The disclosed techniques can be used forsystems with more than 16 SS as well.

Characteristics of Phase Noise

To describe the phase noise in the system:

-   -   Total phase noise PSD is the combination of the PSD at TX and        PSD at RX.    -   Usually UE will have a lot worse PSD than TRP due to the        relative difference in quality of RF chains at UE and TRP.    -   The UEs in the network will be categorized into UE categories        depending on this RF quality and other transmission capabilities        of that UE. That UE category and/or UE capability and/or UE        feature is expected to reflect the PSD quality to some level.

In an OFDM system, the Phase Noise impact is measured on a per symbollevel via:

-   -   CPE (Common Phase Error)    -   ICI (Inter Carrier Interference)

The PSD of Phase Noise is a combination of:

-   -   PSD of PN of Clock    -   PSD of PN of PLL Loop Filter    -   PSD outside the Loop Filter BW

Now, to compensate the PN for OFDM at less than 40 GHz, estimating andcompensating for CPE is sufficient. CPE is different for each symbol,and is a random variable in time, following a type of random walkprocess. The 40 GHz upper bound is an example of current RF technology;however, it should be understood that the upper bound will likely shiftin the future as RF technology progresses.

Measurement of the Phase Noise Between Spatial Streams Descriptors ofPhase Noise Correlation

Depending on the clocks and LOs involved and their mapping to theantenna elements, the Phase Noise process of different SS can becorrelated to different degrees. As such, the following arecharacterized:

-   -   Dependence of the complete phase noise between SS    -   Dependence of the CPE between SS

With respect to dependence of the phase noise between SS, it isdesirable to estimate exact PN samples over time/SC (subcarrier) per SSand then find the cross correlation between the SS of this complete PNprocess. These will include the impact of CPE and ICI.

With respect to dependence of the CPE between SS, there are a number ofoptions to characterize it as provided below. First, CPE is found for asymbol for each SS. Then, the following options can be performed:

-   -   Compare the instantaneous CPE per SS in that symbol:        -   If all SS are completely correlated, then this CPE should be            the same.        -   If there is partial correlation among some SS, but not the            others, then the CPE of those SS should be similar while            that of the uncorrelated SS will be different.    -   Compare the time series of instantaneous CPE per SS over        multiple symbols in the TTI:        -   There are multiple ways to find the CPE in a symbol per SS.        -   If the CPE is estimated over N symbols out of M in the TTI,            then the time series of CPE is obtained per SS.        -   The cross correlation of the CPE over all or some of the SS            can then be calculated.    -   Compare over multiple TTIs:        -   Knowledge from past TTIs in which the UE has been scheduled            can be used to get more sample points for estimating the            CPE.

Methods to Estimate the CPE

It is proposed to use a toolkit of methods to estimate the CPE dependingon the symbol number and structure. They are introduced here with moredetails being provided below.

-   -   DMRS (demodulation-reference signal) based        -   Use the DMRS which is either front loaded or also can be in            middle of the slot    -   PTRS based        -   Use the PTRS in the PDSCH (physical downlink shared channel)    -   CSI-RS (channel state information-reference signal) based        -   The CSI-RS can be used, present again in PDSCH    -   Blind data based        -   The previously presented and patented blind data based            methods can be used in symbols.

Depending on the numerology and frame structure, each symbol in the TTIcan support all or some of the above methods. Using all or some of themethods listed above, one can estimate the CPE time series in a TTI.Then, using one of the CPE cross correlation methods listed above, onecan estimate to what degree the phase noise between the SS iscorrelated.

FIG. 5 provides an example diagram of two subframes 500 transmitted froma MIMO base station that include two orthogonal PTRS ports 502 with thePDSCH 508. The PDCCH (physical downlink control channel) 504 symbols areincluded at the beginning of each frame followed by DMRS (demodulationreference signal) symbols 506 for 12 spatial streams.

Feedback of the Correlation of PN Between SS

The following methods are proposed to feedback the correlation from theUE to the base station. The base station can then use this feedback toallocate PTRS ports to UEs.

Feedback Method Type 1:

Send back the raw or quantized version of the cross correlation matrix.The matrix looks like the matrix below:

SS_xcorr = [C_1_1, C_1_2, …  , C_1_N; C_2_1, C_2_2, C_2_3, …  , C_2_N; …  ; …  ; C_N_1, C_N_2, …  , C_N_N]$\mspace{79mu} {{SS}_{xcorr} = \begin{bmatrix}C_{11} & \ldots & C_{1N} \\\vdots & \ddots & \vdots \\C_{N\; 1} & \; & C_{NN}\end{bmatrix}}$

It is noted that not all values may typically be estimated by a specificUE. It depends on the number of DMRS that a UE can receive. The UE canbe configured to send feedback for cross correlation back to the gNBover a UCI (Uplink scheduling Control Information) message, a MAC CE(Media Access Control-Control Element), a RRC (Radio Resource Control)message, and/or some other desired message channel.

Example

If a TRP transmits SS 1, 2, 3, 4, 5, 6 in one DMRS group (which areusually Quasi Co-Located (QCLed)) and 6 other SS in a second DMRS group;and if UE1 is configured to receive data addressed to it in spatialstreams 1, 2, 3, 4 out of the 12 SS transmitted by that TRP; and if UE1can also receive the other DMRS in the group, i.e. corresponding to SS 5and 6; then the UE1 can use the DMRS to equalize 6 out of the 12 spatialstreams (i.e., SS 1, 2, 3, 4, 5, 6) even though only 4 contain dataaddressed to it. And UE1 can fill in a 6×6 cross correlation matrix.

After this matrix is prepared, it can be signaled back by the UE in anumber of ways.

-   -   Feedback all elements of the matrix: [C_1_1, C_1_2, . . . ,        C_1_N, C_2_1, C_2_2, C_2_3, . . . , C_2_N, . . . , C_N_1, C_N_2,        . . . , C_N_N];    -   Feedback only the off-diagonal terms of the matrix: [C_1_2,        C_1_3, . . . , C_1_N, C_2_1, C_2_3, . . . , C_2_N, . . . ,        C_N_1, C_N_2, . . . , C_N_N−1];    -   Feedback the averaged cross correlation per pair of Spatial        Streams:        -   combine C_1_2 and C_2_1 into Ceff 1_2=(C_1_2+C_2_1)/2        -   send back [Ceff_1_2, Ceff_1_3, Ceff_1_4, . . . , Ceff_1_N,            Ceff_2_3, Ceff_2_4, . . . , Ceff_2_N, Ceff_3_4, . . . ,            Ceff_3_N, Ceff_N−1_N];    -   Feedback only those cross-correlation values which are        above/below a certain threshold;    -   Feedback only those cross-correlation values which are        above/below a certain SS-specific threshold;    -   Feedback quantized (e.g., linear or logarithmic)        cross-correlation values, for example the cross-correlation        value has to be mapped to a two-digit binary number; and/or    -   Consider all the options mentioned above but feedback only those        values that differ from the previous reporting.        It is further noted that additional and/or different feedback        techniques could also be used by the UE to send cross        correlation feedback information back to the TRP, such as a base        station.

Feedback Method Type 2:

In this method, the concept is to send back to the TRP the UE'ssuggestion for allocating the PTRS ports to it. The UE selects theoptimum number of PTRS ports it needs and which DMRS ports with whichthey are associated using the cross correlation matrix it has computed.The raw cross correlation data is not sent back. Instead, the UErecommendation for the PTRS ports it needs is sent back. It is up to thebase station scheduler to take into consideration such feedback from allrelevant scheduled UEs and allocate the actual number of PTRS ports andmap them to the DMRS ports and spatial streams. The feedback can beconfigured to be sent back to the gNB over a UCI message, a MAC CE, aRRC command, and/or some other desired message channel.

Example

Continuing the example from the feedback Type 1 above, the UE measuresthe 6×6 cross correlation matrix, and in this instance finds that thephase noise process on spatial streams 1 and 2 are highly correlated,and that those on spatial streams 3, 4, 5, 6 are highly uncorrelated. Inthat case, it may request the TRP to allocate five spatial streams andto map PTRS port 1 to DMRS port 1, to map no PTRS ports to DMRS port 2,and to map PTRS ports 2 through 5 to DMRS ports 3 through 6,respectively. It is noted that DMRS port 5 and 6 are used for another UEfor this example.

Example Flow Diagrams

FIGS. 6A-C provide an example flow diagram for performing the PNcorrelation at the UEs and for the feedback of these correlations fromthe UE to the base station. It is noted that although the discussionsherein focus on the DL, these methods can be extended to UL in a similarmanner.

Looking first to FIG. 6A, at block 602, when PTRS communications aresupported by the base station, the process starts off with transmittingSlot/TTI with 1 PTRS per DMRS port with some known time frequencydensity. Otherwise, there is lesser PTRS per DMRS ports according tosome implicitly/explicitly indicated association rule. Further:

-   -   There is one DMRS port per spatial stream;    -   The spatial streams can be transmitted from any of the setups        indicated: Multi TRP, One TRP, to single UE, to multiple UEs;        and    -   Time frequency density/location can be implicitly/explicitly        indicated to the UE.

Next, at block 604, at each UE scheduled in that TTI, the following aredone in the front loaded DMRS symbol:

-   -   Estimate CPE per Spatial Stream using all the DMRS in the        scheduled resource blocks for that UE; and    -   DMRS from spatial streams not meant for the given UE group or        DMRS group are also used if possible (this is optional). If not,        limit the CPE estimation to SS meant for that UE group or DMRS        group only

Next, at block 606, at each UE scheduled in that TTI, the following aredone in the symbols that have PTRS (special case is if every PDSCHsymbol has a PTRS):

-   -   Estimate CPE per Spatial Stream using all the PTRS in the        scheduled resource blocks for that UE; and    -   PTRS from spatial streams not meant for the given UE are also        used if possible (this is optional). If not, limit the CPE        estimation to use PTRS in SS meant for that UE only.

Next, at block 608, at each UE scheduled in that TTI, if additional(non-front loaded) DMRS symbol present, the following are done:

-   -   Estimate CPE per Spatial Stream using all the DMRS in the        scheduled resource blocks for that UE; and    -   DMRS from spatial streams not meant for the given UE group or        DMRS group are also used if possible (this is optional). If not,        limit the CPE estimates to SS meant for that UE group or DMRS        group only

Next, at block 609, if some of the symbols in the TTI contain a CSI-RStransmission, the UE can use the CSI-RS to derive the CPE if sufficientnumber of CSI-RS are available to achieve reliable CPE estimation.

Next, at block 611, at each UE scheduled in that TTI, if a selectedPDSCH symbol has no DMRS or PTRS or CSI-RS, then estimate the CPE withblind method. This is done only in SS meant for that UE.

Next, at block 612, the cross correlation of the CPE is calculatedacross the spatial streams. This is done for as many SS as possible.

If the Type 1 Feedback method is being used, at block 614, the UE thentransmits the complete N×N Cross Correlation matrix, or one of thecondensed and transformed version of it, back to the base station.

If the Type 2 Feedback method is being used, at block 616, the UE thenselects the optimum number of PTRS ports it needs and which DMRS portswith which they are associated. This recommended PTRS port list istransmitted back to the base station(s) by the UE.

Finally, at block 618, the base stations (e.g., gNB) updates PTRSallocation and mapping to DMRS ports based on reports received from UEs.

Extension/Modification for Multi TTI Operation

The following provides an extension of the example flow provided inFIGS. 6A-C for OFDM to multiple TTIs. The proposed method uses the CPEestimates from the last M number of TTIs for which a specific UE wasscheduled with a similar transmission mode setup.

The proposed method for each TTI-i is as follows:

-   -   If the UE is scheduled with Transmission scheme setup Y:        -   Estimate CPE in each possible symbol of the TTI as provided            in the example of FIGS. 6A-C; and        -   Obtain Time Series of CPE per symbol per spatial stream.    -   If previous TTI(s) that used same Transmission scheme Y were        present:        -   Concatenate the CPE Estimates of Current TTI-i with the past            TTI(s) and update the Cross Correlation Matrix (SS_xcorr);            -   OR        -   Find SS_xcorr per individual TTI and combine them with            either averaging or a variety of exponentially weighted            averaging to get a new update of SS_xcorr.    -   Send back the updated SS_xcorr or the recommended PTRS port        configuration settings back to the gNB depending on the Feedback        Method Type 1 or 2 used.    -   Wait for next TTI when the UE is scheduled.

Extension/Modification for Single Carrier Modulations

For the case of single carrier waveforms [SC=Single Carrier TransmissionScheme], such as Null CP Single Carrier Waveform, or SC-FDMA or any ofthe other candidates in the single carrier family, the problem of phasenoise estimation and compensation still exists. However, the OFDM typeCommon Phase Error (CPE) is not relevant as the phase noise does notneed to be compensated in frequency (per subcarrier) and instead needsto be compensated in time.

Some techniques for such PN time compensation are:

-   -   Time domain Pilot aided compensation;    -   Time domain Cyclic Prefix aided compensation; and    -   Time domain blind compensation.

In addition, the proposed techniques described herein can be extended tosuch single carrier scenario. For example, consider a MU-MIMO type SCsystem where multiple UEs are scheduled on the same time frequencyresource. Each UE gets a UE specific pilot that is pre-coded in the samemanner as the data to the UE. This is called the SC DMRS pilot. Thereare additional SC PTRS pilots that can be used in the system. Each SCPTRS is associated to one or more SC DMRS port and share the sameprecoding as exactly one SC DMRS port. The algorithm can then beimplemented the same as in FIGS. 6A-C above for the OFDM case but PTRSis replaced by SC PTRS and DMRS is replaced by SC DMRS.

As noted above, although the discussions herein focus on the DL, thesemethods can be extended to UL in a similar manner.

It is noted that the disclosed embodiments can be used with respect to avariety of OFDM-based transmission schemes for RF communication systems.It is also noted that as used herein, a “radio frequency” or RFcommunications means an electrical and/or electro-magnetic signalconveying useful information and having a frequency from about 3kilohertz (kHz) to thousands of gigahertz (GHz) regardless of the mediumthrough which such signal is conveyed. The OFDM-based transmissions maybe transmitted through a variety of mediums (e.g., air, free space,coaxial cable, optical fibers, copper wire, metal layers, and/or otherRF transmission mediums). As one example, the disclosed embodimentscould be used for millimeter (mm) wave transmissions between 30-300 GHzhaving wavelengths of 1-10 mm (e.g., a transmission range of 71-76 GHz)if OFDM-based modulation were used for the mm wave transmissions. Inaddition, the disclosed embodiments will likely be useful for 5Gsolutions up to 40 GHz where OFDM-based modulations are more likely tobe implemented. For example, 5G frequency ranges and bands around 28GHz, 39 GHz, and/or other frequency ranges or bands where OFDM-basedmodulation is used for RF transmissions will benefit from the disclosedtechniques. It is further noted that example wireless communicationsystems within which the disclosed techniques can be applied are alsodescribed in U.S. Published Patent Application No. 2015-0303936 (Ser.No. 14/257,944) and U.S. Published Patent Application No. 2015-0305029(Ser. No. 14/691,339), each of which is hereby incorporated by referencein its entirety.

FIG. 7 is a block diagram of an example embodiment for a communicationsystem 700 that can transmit and receive OFDM symbols as describedherein. The example embodiment of FIG. 7 includes a transmit path and areceive path. The transmit path includes multi-FPGA processing circuitry702, a baseband transmitter 704, an IF upconverter 706, and an RFtransmitter 708. The receive path includes an RF receiver 712, an IFdownconverter 714, a baseband receiver 716, and multi-FGPA processingcircuitry 718. The transmit path and the receive path that arecommunicating with each other can be located in different devices (e.g.,base station and user equipment for cellular communications). Ifbi-directional communications are desired, the different devices caneach include a transmit path and a receive path. Other variations canalso be implemented.

Looking to the transmit path, transmit data 722 is sent to multipleFPGAs 702 that provide multi-FPGA processing of the transmit data 722.The transmit data 722 can be generated by other processing circuitrysuch as a control processor or other circuitry. These FPGAs 702 canoperate at a selected clock rate (e.g., 192 MS/s (mega samples persecond) or other rate) and can use efficient parallel wide data pathimplementations, for example, with multiple (e.g. 16) data elements(e.g., baseband samples) per wide data path sample. The FPGAs 702 outputdigital baseband signals 724 to the baseband transmitter 704. Thebaseband transmitter 704 includes a digital-to-analog converter (DAC)that converts the digital baseband samples to analog baseband signals726. The baseband transmitter 704 including the DAC can operate at aselected sampling rate (e.g., 3.072 GS/s (Giga samples per second) orother rate) and can receive digital baseband samples from one ormultiple FPGAs 702 within the multi-FPGA processing circuitry. Theanalog baseband signals 726 are received by an IF (intermediatefrequency) upconverter 706 that mixes the analog baseband signals 726 tohigher frequency IF signals 728. These IF signals 728 are received bythe RF transmitter 708 which further upconverts these signals to thefrequency range of the desired transmissions.

Looking to the receive path, the receiver 712 receives the RFtransmissions from the RF transmitter which can be within a desiredfrequency range. The RF receiver 712 downconverts these RF transmissionsto lower frequency IF signals 732. The IF signals 732 are then receivedby an IF downconverter 714 that mixes the IF signals 732 down to analogbaseband signals 734. The analog baseband signals 723 are then receivedby a baseband receiver 716. The baseband receiver 716 includes ananalog-to-digital converter (ADC) that converts the analog basebandsignals 734 to digital baseband signals 736. The baseband receiver 716including the ADC can operate at a sampling rate (e.g., 3.072 GS/s (Gigasamples per second) or other rate) and can send digital baseband samples736 to one or multiple FPGAs within the multi-FPGA processing circuitry718. The FPGAs 718 receive the digital baseband signals 736 and generatedigital data that can be processed by additional processing circuitrysuch as a control processor or other circuitry. These FPGAs 718 canoperate at a selected rate (e.g., 192 MS/s (mega samples per second) orother rate using efficient parallel wide data path implementations, forexample, with multiple (e.g., 16) data elements (e.g., baseband samples)per wide data path sample.

FIG. 8 is a block diagram of an example embodiment for circuitry thatcan be used to provide a baseband receiver, a baseband transmitter,multi-FPGA processing circuitry, and a real-time processor with respectto the embodiment of FIG. 7. The embodiment of FIG. 8 provides tworeceive/transmit streams and related processing circuitry.

Looking to the embodiment of FIG. 8, two analog-to-digital converters(ADC1 802-1, ADC2 802-2) receive analog baseband signals and outputsampled digital baseband signals 804 (i.e., sampled time-domain basebandreceive signals) to two demodulators/equalizers 806. Thedemodulators/equalizers (DEMODULATOR/EQUALIZER 1 806-1,DEMODULATOR/EQUALIZER 2 806-2) demodulate and equalize the respectivereceive signals. Due to the complexity of the MIMO (multiple inputmultiple output) equalization task, some parts of the relatedfunctionality are realized by a separate MIMO processing circuitry (MIMOPROCESSING 808). Specifically, this MIMO processing circuitry 808performs the MIMO channel estimation and the calculation of theequalizer weights 812. For this, it uses the (pre-processed) pilotsignals/symbols 814 extracted from both received baseband signals asinput. These (pre-processed) pilot signals 814 are provided by thedemodulators/equalizers 806. The equalizer weights 812 (W1, W2)calculated by the MIMO processing circuitry 808 are fed back to thedemodulators/equalizers 806, which can perform the final MIMOequalization using these equalizer weights 812. To support this finalMIMO equalization task, the demodulators/equalizers 806 can exchangeintermediate equalization results. The final output of thedemodulators/equalizers 806 are equalized QAM (quadrature amplitudemodulation) symbols 816 for both receive streams. These equalized QAMsymbols 816 are provided to the MIMO processing circuitry 808, which candistribute the equalized QAM symbols 824 to multiple decoders (DECODER822). It is noted that the upper set of decoders 822 can be used fordecoding the first receive stream and the lower set of decoders 822 canbe used for decoding the second stream. The decoders 822 output decodeddigital receive data 828 plus CRC (cyclic redundancy check) results pertransport block to the MAC support FPGA 826. The MAC support FPGA 826can collect the output data 828 of all decoders 822, can further processthem, and can provide them to the real-time processor (REAL-TIMEPROCESSOR 832) in a synchronized and consistent manner. The real-timeprocessor 832 can perform further operations on the received data 828(and CRC results) provided by the MAC support FPGA 826. Further, it canprovide receiver (RX) control information 836 to the MAC support FPGA826 and/or other receiver FPGAs (not shown) to control and configure therespective receivers. For example, the real-time controller 832 canprovide the control data 836 for all decoders 822 per subframe to theMAC support FPGA 826, and the MAC support FPGA 826 can distribute thesecontrol data 838 to each decoder 822 to provide the configuration usedto decode the related transport block. A similar functionality can beprovided by the real-time processor 832 for the transmit paths. Uncodeddigital transmit data 842 and related transmitter (TX) control data 844are sent from the real-time processor 832 to the MAC support FPGA 826,which distributes the digital transmit data 846 as well as therespective encoder/modulator control data 848 to the twomodulators/encoders (MODULATOR/ENCODER 1 852-1, MODULATOR/ENCODER 2852-2). The modulators/encoders 852 encode the transmit data 846 andperform the transmit modulation, e.g., generate the digital time-domainbaseband transmit signals 854. These digital time-domain basebandtransmit signals 854 are sent by the modulators/encoders 852 to thedigital-to-analog converters (DAC1 856-1, DAC2 856-2). The DACs 856receive the digital baseband signals 854 and output analog basebandsignals. It is noted that demodulators/equalizers, modulators/encoders,MIMO processing circuitry, and decoders can be implemented usingmultiple parallel FPGAs.

The disclosed embodiments can also be used for OFDM-based transmissionschemes for massive MIMO cellular telecommunication systems as describedin U.S. Published Patent Application 2015/0326291, entitled “SIGNALINGAND FRAME STRUCTURE FOR MASSIVE MIMO CELLULAR TELECOMMUNICATIONSYSTEMS,” which is hereby incorporated by reference in its entirety.Such massive MIMO (multiple input, multiple output) communicationsystems can be used for 5G dynamic TDD (time division duplex) airinterfaces. The 5G (5th generation) mobile telecommunications is able tospan a wide variety of deployment scenarios (e.g., Rural, Urban Macro,Dense Urban, Indoor, etc.) in a flexible and scalable manner. Inparticular, massive MIMO reciprocity-based TDD air interfaces allow forsymbol-level switching and potential configurability that in turn allowfor features to support three primary aspects of 5G air interfaces,namely enhanced Mobile BroadBand (eMBB), massive Machine TypeCommunications (mMTC) and Ultra-Reliable and Low Latency Communications(URLLC).

The disclosed embodiments can also be used with CPE compensationtechniques and related embodiments described in FIGS. 10 through 15 andin U.S. patent application Ser. No. 15/855,148, filed Dec. 27, 2017,which claims priority to U.S. Provisional Patent Application Ser. No.62/443,226, entitled “BLIND COMMON PHASE ERROR (CPE) COMPENSATION FOROFDM SYMBOLS IN WIRELESS COMMUNICATION SYSTEMS,” each of which is herebyincorporated by reference in its entirety.

FIG. 9 is a diagram of an example embodiment 900 for electroniccomponents that can be used to implement a base station and/or userequipment (UE) including the functions and operational featuresdescribed for the disclosed embodiments. For the example embodiment 900shown in FIG. 9, one or more processors 908 communicate with othercomponents through system bus interconnect 902. For example, the one ormore processors 908 communicate with input/output (I/O) circuitry 904and transmit/receive (TX/RX) circuitry 906 through the system businterconnect 902. Additional circuitry can also be included such aspower supply circuitry and/or other desired circuitry. The TX/RXcircuitry 906 provides one or more cellular radios and are preferablycoupled to a plurality of antennas through which the TX/RX circuitrytransmits and receives RF (radio frequency) signals (e.g., from a fewkHz to 10 GHz and above). The I/O circuitry 904 provides one or moreinterfaces for users, such as graphical user interfaces, and/orconnections to peripheral devices (e.g., displays, keyboards, mice,point device, and/or other I/O peripheral devices). The memory 910 isalso coupled to the system bus interconnect 902 and can be used by theone or more processors 908 to load and/or store instructions, data,and/or other information during operation. One or more data storagedevice(s) 912 are also connected to the system bus interconnect 902 andcan store software or program instructions and/or other desired data orinformation for the operation of the processing system. For example,computer-readable instructions stored in the data storage devices 912can be loaded within the memory 910 and then executed by theprocessor(s) 908 to carry out the functions described herein.

It is noted that different and/or additional components from thosedepicted in FIG. 9 could also be used to implement one or more radiosystems for the embodiments described herein while still takingadvantage of the techniques described herein. It is further noted thatthe system bus interconnect 902 can be implemented as multipleinterconnection buses with our without additional intervening circuitrysuch as routing or switching circuitry. Further, the processor(s) 908can be implemented using one or more programmable integrated circuitsincluding controllers, microcontrollers, microprocessors, hardwareaccelerators, configurable logic devices (e.g., field programmable gatearrays), and/or other programmable integrated circuits that areprogrammed to carry out the function described herein. Further, the oneor more processor(s) 908 can execute instructions stored in anon-transitory tangible computer-readable medium to perform thefunctions described herein. In addition, data storage device(s) 912 canbe implemented as any desired non-transitory tangible medium that storesdata, such as data storage devices, FLASH memory, random access memory,read only memory, programmable memory devices, reprogrammable storagedevices, hard drives, floppy disks, DVDs, CD-ROMs, and/or any othernon-transitory data storage mediums. The memory 910 can be any datastorage medium configured to maintain data storage when powered. Othervariations could also be implemented.

It is also noted that the functional blocks described herein can beimplemented using hardware, software, or a combination of hardware andsoftware, as desired. In addition, one or more processors or processingcircuitry running software and/or firmware can also be used, as desired,to implement the disclosed embodiments. It is further understood thatone or more of the operations, tasks, functions, or methodologiesdescribed herein may be implemented, for example, as software orfirmware and/or other program instructions that are embodied in one ormore non-transitory tangible computer readable mediums (e.g., memory)and that are executed by one or more controllers, microcontrollers,microprocessors, hardware accelerators, and/or other processors orprocessing circuitry to perform the operations and functions describedherein.

It is further noted that the functional blocks, devices, and/orcircuitry described herein can be implemented using hardware, software,or a combination of hardware and software. In addition, one or moreprocessors (e.g., central processing units (CPUs), controllers,microcontrollers, microprocessors, hardware accelerators, programmableintegrated circuitry, FPGAs (field programmable gate arrays), ASICs(application specific integrated circuits), and/or other programmableprocessing circuitry) can be programmed to perform the operations,tasks, functions, or actions described herein for the disclosedembodiments. For example, the one or more electronic circuits can beconfigured to execute or otherwise be programmed with software,firmware, logic, and/or other program instructions stored in one or morenon-transitory tangible computer-readable mediums (e.g., data storagedevices, flash memory, random access memory, read only memory,programmable memory devices, reprogrammable storage devices, harddrives, floppy disks, DVDs, CD-ROMs, and/or any other tangible datastorage medium) to perform the operations, tasks, functions, or actionsdescribed herein for the disclosed embodiments.

It is still further noted that the functional blocks, components,systems, devices, and/or circuitry described herein can be implementedusing hardware, software, or a combination of hardware and software. Forexample, the disclosed embodiments can be implemented using one or moreprogrammable integrated circuits that are programmed to perform thefunctions, tasks, methods, actions, and/or other operational featuresdescribed herein for the disclosed embodiments. The one or moreprogrammable integrated circuits can include, for example, one or moreprocessors and/or PLDs (programmable logic devices). The one or moreprocessors can be, for example, one or more central processing units(CPUs), controllers, microcontrollers, microprocessors, hardwareaccelerators, ASICs (application specific integrated circuit), and/orother integrated processing devices. The one or more PLDs can be, forexample, one or more CPLDs (complex programmable logic devices), FPGAs(field programmable gate arrays), PLAs (programmable logic array),reconfigurable logic circuits, and/or other integrated logic devices.Further, the programmable integrated circuits, including the one or moreprocessors, can be configured to execute software, firmware, code,and/or other program instructions that are embodied in one or morenon-transitory tangible computer-readable mediums to perform thefunctions, tasks, methods, actions, and/or other operational featuresdescribed herein for the disclosed embodiments. The programmableintegrated circuits, including the one or more PLDs, can also beprogrammed using logic code, logic definitions, hardware descriptionlanguages, configuration files, and/or other logic instructions that areembodied in one or more non-transitory tangible computer-readablemediums to perform the functions, tasks, methods, actions, and/or otheroperational features described herein for the disclosed embodiments. Inaddition, the one or more non-transitory tangible computer-readablemediums can include, for example, one or more data storage devices,memory devices, flash memories, random access memories, read onlymemories, programmable memory devices, reprogrammable storage devices,hard drives, floppy disks, DVDs, CD-ROMs, and/or any othernon-transitory tangible computer-readable mediums. Other variations canalso be implemented while still taking advantage of the techniquesdescribed herein.

Looking now to FIG. 10, a block diagram is provided of an exampleembodiment 200 for symbol processing for OFDM transmissions. Atime-frequency synchronization processor 202 receives incoming symbols201 from OFDM transmissions and aligns the start of each OFDM symbol 201for the FFT (fast Fourier transform) operations. The FFT OFDMdemodulator 204 receives the output of the time-frequencysynchronization processor 202 and demodulates the OFDM symbol to extractsubcarriers using an FFT operation. The channel estimator 208 receivesthe subcarriers and uses pilot information within the subcarriers togenerate an estimate of the channel response. The equalization processor206 receives the channel estimate from the channel estimator 208 andapplies it to the extracted subcarriers from the OFDM demodulator 204 togenerate equalized OFDM subcarriers. The CPE estimator 212 receives theequalized OFDM subcarriers and applies one or more blind algorithms 214(e.g., pilot-aided blind method, blind-only method) to generate a CPEestimate, although pilot only estimation can also be applied. The CPEcompensation processor 210 then receives the CPE estimate from the CPEestimator 212 and applies it to the equalized OFDM subcarriers tocompensate for the CPE. The compensated OFDM subcarriers are thendemodulated by the demodulator 216 to generate demodulated data 209. Forexample, this demodulation can produce a decision regarding whichconstellation point was transmitted within a modulation scheme (e.g., 16QAM) applied to the transmitted symbols for processing.

FIG. 11 is a block diagram of an example embodiment 300 where the CPEestimator 212 includes multiple estimation algorithms including a pilotonly CPE estimation algorithm 304, a blind only CPE estimation algorithm214A, and a pilot aided blind CPE estimation algorithm 214B. The CPEestimator 212 receives a control signal 302 that determines which CPEestimation algorithm is applied to any particular OFDM symbol. Thiscontrol signal 302, for example, can be generated by one or more controlprocessors. For the embodiment, the CPE compensator 210 includes adigital mixer 306 that mixes the CPE estimate 308 from the CPE estimator212 with a compensated, or de-rotated, version 315 of the equalized OFDMsubcarriers 312 to generate the CPE compensated OFDM subcarriers 314.The CPE estimate 308 is provided to an accumulator 317 that accumulatesthe CPE estimate 308 with the output 313 of a selector 307 to produce anaccumulated CPE estimate 303, which is provided to a delay block 319that delays the accumulated CPE estimate 303 by one OFDM symbol. Thus,the delay block 319 produces an accumulated CPE estimate 309 associatedwith the previous OFDM symbol, whereas the accumulator 317 produces anaccumulated CPE estimate 303 associated with the current OFDM symbol.The delayed accumulated CPE estimate 309 is provided to an input of theselector 307 and a zero value is provided to the other input of theselector 307. The control signal 302 controls the selector 307 to selectthe delayed CPE estimate 309 when the current OFDM symbol is absentpilot symbols for estimating CPE and selects the zero input when pilotsymbols for estimating CPE are present, or embedded, in the current OFDMsymbol, as well as when the system is at rest and in response to thereceiver transitioning to the blind-only method 214A from the pilot-onlymethod 304 or pilot-aided method 214B. The delayed accumulated CPEestimate 309 is also provided to a second mixer 311 that mixes theequalized subcarriers 312 of the current OFDM symbol to generate thecompensated version 315 of the equalized OFDM subcarriers 312. Thecompensated version 315 of the equalized OFDM subcarriers 312 are alsoprovided to the CPE estimator 212, which uses them to compute the CPEestimate 308. Preferably, the digital mixer 306 forms a unitaryamplitude complex value having a phase, or offset angle, that is anegated version of the CPE estimate 308 and multiplies the compensatedversion of the equalized OFDM subcarriers 315 by the formed complexvalue to generate the CPE compensated OFDM subcarriers 314. Similarly,the second mixer 311 forms a unitary amplitude complex value having aphase, or offset angle, that is a negated version of the delayedaccumulated CPE estimate 309 and multiplies the equalized OFDMsubcarriers 312 by the formed complex value to generate the compensatedversion 315 of the equalized OFDM subcarriers 312. As described above,the CPE estimator 212 uses the de-rotated/compensated equalizedsubcarriers 315 to generate the CPE estimate 308. For example, in thecase of an N-subcarrier FFT, the CPE estimator 212 uses N subcarriers togenerate the CPE estimate 308, and the mixer 306 applies the CPEestimate 308 to the N subcarriers. It is noted that in the case of anOFDM symbol embedded with pilot symbols that are used to compute the CPEestimate 308, the mixer 311 will not modify the equalized OFDMsubcarriers 312 (i.e., will mix them with a unitary value by operationof the selector 307 to output a zero-valued phase, or angle, that, asthe exponent of the formed complex value, will cause it to be unitary).

It is noted that the methods and related systems are provided that adaptthe density of the PN reference signals or pilots within the OFDMtransmissions in a dynamic and/or semi-static manner based on theperformance of the purely pilot aided CPE compensation method 304, blindCPE compensation method 214A, and/or the pilot aided blind CPEcompensation method 214B. This density of PN reference signals or pilotscan be adapted in time and/or frequency. In addition, a receiving device(e.g., one or more UEs) can send control messages back to a transmittingdevice (e.g., one or more base stations) indicating the performancelevel associated with the CPE compensation methods being employed. Forexample, the number of symbols within a duration including PN referencesignals or pilots can be reduced by the transmitting device as long asthe pilot aided CPE compensation method continues to provide adequateperformance. Once all PN reference signals are removed, the blind CPEcompensation method 214A can be used as long as it continues to provideadequate performance. Other variations could also be implemented whilestill taking advantage of the blind CPE estimation techniques describedherein.

FIG. 12 is a process flow diagram of an example embodiment 400 for ablind only CPE estimation algorithm for the CPE estimator 212. As afully blind method, no pilots are necessary for embodiment 400 and nopilot overhead is required. For every symbol, the phase noise isestimated using a blind algorithm. It is noted that one or more ofvarious PN estimators can be used for the blind CPE estimation. For theexample embodiment 400, a threshold and average estimation method isused where received IQ points are thresholded and within eachthresholding region, the phase of the I and Q components are averaged toderive the CPE estimate. A power law PN estimation method could also beused where received IQ points are raised to their Mth power, and theresult is averaged and then post processed to derive the CPE estimate.Other blind PN estimation methods could also be used without requiringPN reference signals or other pilots to provide CPE estimation. Anexample of a fourth-power law estimation method that may be used in aQAM constellation that has quadrant symmetry (i.e., is symmetric withrespect to phase

$\frac{\pi}{2},$

e.g., square or cross-QAM constellations) is shown in equation (1)below.

$\begin{matrix}{\theta = {\frac{1}{4}\mspace{14mu} {{angle}\mspace{14mu}\left\lbrack {{E\left( {X^{*4}(n)} \right)}\frac{\sum\limits_{n = 1}^{N}{Y^{4}(n)}}{N}} \right\rbrack}}} & (1)\end{matrix}$

In equation (1), θ is the CPE estimate, E is the expectation operator,X(n) are the values of the known transmitted QAM constellation signalset, the * operator denotes the complex conjugate of the value (in thiscase, the subcarrier), Y(n) are the received subcarriers of the OFDMsymbol, and N is the OFDM symbol size, i.e., the number of subcarriers.

Looking in more detail to FIG. 12, the initial CPE estimate is initiallyset to zero in block 402. In block 404 (STEP 2), a set of decisionregions is defined. For example, a set of eight regions of a 16-QAMmodulation constellation may be defined, as shown in FIG. 14 as 602,604, 606, 608, 612, 614, 616 and 618, as described below in more detail.In block 406 (STEP 3), data is extracted for all data subcarriers withinone of the regions defined in block 404 (STEP 2). For example, all thedata subcarriers with a real (I) magnitude within region 602 may beextracted, e.g., I≤−2/sqrt(10), where “sqrt” is a square root function.In block 408 (STEP 4), the CPE is estimated on the set of datapreviously extracted in block 406 (STEP 3) to obtain a partial CPEestimate. For example, a linear fitting algorithm can be applied to theextracted data, as shown in FIG. 15. Examples of linear fitting schemesinclude least square estimation, maximum likelihood, Bayesian linearregression, and other linear fitting schemes. In block 410 (STEP 5), theprocessing in blocks 406 and 408 (STEPS 3 and 4) are repeated to coverall remaining regions defined in block 404 (STEP 2) to obtain a partialCPE estimate for each region. In block 412 (STEP 6), all the partial CPEestimates from the threshold regions obtained according to blocks 406,408 and 410 (STEPS 3, 4 and 5) are combined, for example by averaging,to generate a final blind CPE estimate. As noted below, differentthreshold values and regions could also be used, and the number ofthreshold regions could also be adjusted.

FIG. 13 is a process flow diagram of an example embodiment 500 for apilot aided blind CPE estimation algorithm for the CPE estimator 212.This embodiment provides a combination of the fully blind method abovewith traditional pilot aided CPE estimation methods. One traditionalapproach to reduce the PN pilot overhead for pilot aided CPE estimationis to have pilots only on intermittent symbols, for example on everysecond OFDM symbol. In embodiment 400, the pilot aided CPE estimate isstarted as the baseline. The blind estimator is then run on the symbolsthat do not have the pilots embedded in them.

Looking in more detail to FIG. 13, it is assumed that a traditionalpilot only CPE estimation has already been run. In block 502 (STEP 1),the initial CPE estimate is set to the result of the pilot only basedCPE estimate of the most recent symbol with an embedded PN referencesignal or pilot. In block 504 (STEP 2), blocks 404, 406, 408, and 410(STEPS 2-5) of embodiment 400 for fully blind CPE estimation areperformed for all subsequent symbols without PN pilots. In block 506(STEP 3), the initial pilot CPE estimate from block 502 (STEP 1) iscombined with the blind CPE estimate from block 504 (STEP 2) to generatefinal pilot aided blind CPE estimate. The initial pilot CPE estimate maybe combined with the blind CPE estimate by accumulating the twoestimates and continuing to accumulate succeeding estimates, as shown inthe embodiment of FIG. 11, for example.

FIG. 14 is a diagram of an example embodiment 600 for decisionthresholds for the different 16-QAM threshold regions described abovewith respect to FIG. 12. The x-axis represents the real (I) magnitudesfor the modulation scheme, and the y-axis represents the imaginary (Q)magnitudes for the modulation scheme. For the real (I) thresholdsregions, region 602 includes values for I such that I≤−2/sqrt(10);region 604 includes values for I such that −2/sqrt(10)<I≤0; region 606includes values for I such that 0<I≤2/sqrt(10); and region 608 includesvalues for I such that I>2/sqrt(10). For the imaginary (Q) thresholdsregions, region 612 includes values for Q such that Q≤−2/sqrt(10);region 614 includes values for Q such that −2/sqrt(10)<Q≤0; region 616includes values for Q such that 0<Q≤2/sqrt(10); and region 618 includesvalues for Q such that Q>2/sqrt(10). It is noted that differentthreshold values and regions could also be used, and the number ofthreshold regions could also be adjusted. For example, for 64-QAMmodulation 16 different threshold regions could be used, and for QPSKmodulation four different threshold regions could be used, respectively.Other variations could also be implemented while still taking advantageof the blind CPE estimation techniques described herein.

It is noted that the disclosed embodiments can be used with respect to avariety of OFDM-based transmission schemes for RF communication systems.It is also noted that as used herein, a “radio frequency” or RFcommunications means an electrical and/or electro-magnetic signalconveying useful information and having a frequency from about 3kilohertz (kHz) to thousands of gigahertz (GHz) regardless of the mediumthrough which such signal is conveyed. The OFDM-based transmissions maybe transmitted through a variety of mediums (e.g., air, free space,coaxial cable, optical fibers, copper wire, metal layers, and/or otherRF transmission mediums). As one example, the disclosed embodimentscould be used for millimeter (mm) wave transmissions between 30-300 GHzhaving wavelengths of 1-10 mm (e.g., a transmission range of 71-76 GHz)if OFDM-based modulation were used for the mm wave transmissions. Inaddition, the disclosed embodiments will likely be useful for 5Gsolutions up to 40 GHz where OFDM-based modulations are more likely tobe implemented. For example, 5G frequency ranges and bands around 28GHz, 39 GHz, and/or other frequency ranges or bands where OFDM-basedmodulation is used for RF transmissions will benefit from the blind CPEcompensation techniques described herein for the disclosed embodiments.It is further noted that example wireless communication systems withinwhich the disclosed blind CPE compensation techniques can be applied arealso described in U.S. Published Patent Application No. 2015-0303936(Ser. No. 14/257,944) and U.S. Published Patent Application No.2015-0305029 (Ser. No. 14/691,339), each of which is hereby incorporatedby reference in its entirety.

FIG. 15 is a diagram of an example embodiment 1000 employing the 8decision thresholds for the different 16-QAM threshold regions of FIG.14 and employing the embodiment of FIG. 12 method to extract subsets ofdata subcarriers of an OFDM symbol that fall within the 8 regions, i.e.,the four real part regions 602, 604, 606 and 608, and the four imaginarypart regions 612, 614, 616 and 618. FIG. 15 further illustrates fittinga line to the extracted subcarriers that fall within region 602 (one ofthe four vertical real part regions) and computing the angle between thefitted line and a phase noise absence line (e.g., a vertical line) toobtain the region 602-specific CPE estimate. FIG. 15 further illustratesfitting a line to the extracted subcarriers that fall within region 614(one of the four horizontal imaginary part regions) and computing theangle between the fitted line and a phase noise absence line (e.g., ahorizontal line) to obtain the region 614-specific CPE estimate. Similarsubcarrier extractions, line fitting, and CPE estimate calculations maybe performed for additional regions, as necessary, and then theregion-specific CPE estimates can be averaged to obtain a final, oroverall, CPE estimate for the OFDM symbol, e.g., according to FIG. 12.As described above, the estimated angle may form the phase of a unitaryamplitude complex number to be multiplied by each subcarrier of an OFDMsymbol to accomplish compensation of the CPE in the OFDM symbolsubcarriers. It is noted that the extracted subcarriers upon which thelines are fit are compensated/de-rotated subcarriers (e.g., output 315of mixer 311 of FIG. 11), which may advantageously result in a moreaccurate CPE estimate 308 by decreasing the likelihood that theindividual subcarriers will be extracted into the wrong region. Asdescribed above, advantageously the number of regions for whichregion-specific CPE estimates may be computed and averaged may varybased on available computing power and needed CPE compensationefficiency.

Further modifications and alternative embodiments of this invention willbe apparent to those skilled in the art in view of this description. Itwill be recognized, therefore, that the present invention is not limitedby these example arrangements. Accordingly, this description is to beconstrued as illustrative only and is for the purpose of teaching thoseskilled in the art the manner of carrying out the invention. It is to beunderstood that the forms of the invention herein shown and describedare to be taken as the presently preferred embodiments. Various changesmay be made in the implementations and architectures. For example,equivalent elements may be substituted for those illustrated anddescribed herein, and certain features of the invention may be utilizedindependently of the use of other features, all as would be apparent toone skilled in the art after having the benefit of this description ofthe invention.

1. A method, comprising: receiving, by a first wireless base station oruser equipment (BS or UE), a frequency division multiplexed (FDM) symboltransmitted by a second wireless BS or UE; processing, by the first BSor UE, the received FDM symbol to obtain its equalized FDM datasubcarriers; generating, by the first BS or UE, a CPE estimate using theequalized FDM data subcarriers; sending, by the first wireless BS or UEto the second wireless BS or UE, control messages that indicate a CPEcompensation performance level using the CPE estimate; and wherein thesecond wireless BS or UE, in response to the control messages, isenabled to adapt a density in time and/or frequency of embedded pilotsymbols within FDM symbols subsequently transmitted to the firstwireless BS or UE.
 2. The method of claim 1, wherein to adapt thedensity of embedded pilot symbols, the second wireless BS or UE reducesthe density of embedded pilot symbols in the subsequently transmittedFDM symbols as long as the control messages continue to indicate anadequate CPE compensation performance level.
 3. The method of claim 1,further comprising: using, by the first wireless BS or UE, a blind CPEcompensation method for subsequently received FDM symbols that arewithout embedded pilot symbols as long as the control messages continueto indicate an adequate CPE compensation performance level.
 4. Themethod of claim 1, further comprising: compensating, by the first BS orUE, each of the equalized FDM data subcarriers using the CPE estimate.5. The method of claim 1, wherein the second wireless BS or UE adaptsthe density of embedded pilot symbols in a dynamic manner.
 6. The methodof claim 1, wherein the second wireless BS or UE adapts the density ofembedded pilot symbols in a semi-static manner.
 7. The method of claim1, wherein said generating a CPE estimate comprises generating a CPEestimate based on a constellation diagram associated with a modulationscheme used by the second wireless BS or UE to generate the FDM datasubcarriers of the transmitted FDM symbol.
 8. The method of claim 1,wherein the control messages that indicate the CPE compensationperformance level comprise RRC control messages.
 9. A wireless basestation or user equipment (BS or UE), comprising: a receiver, configuredto receive a frequency division multiplexed (FDM) symbol transmitted byanother wireless BS or UE; and a processor, configured to: process thereceived FDM symbol to obtain its equalized FDM data subcarriers;generate a CPE estimate using the equalized FDM data subcarriers; andsend to the other wireless BS or UE control messages that indicate a CPEcompensation performance level using the CPE estimate; and wherein theother wireless BS or UE is enabled by the control messages to adapt adensity in time and/or frequency of embedded pilot symbols within FDMsymbols subsequently transmitted to the wireless BS or UE.
 10. Thewireless BS or UE of claim 9, wherein the other wireless BS or UE isenabled by the control messages to adapt the density of the embeddedpilot symbols by reducing the density of embedded pilot symbols in thesubsequently transmitted FDM symbols as long as the control messagescontinue to indicate an adequate CPE compensation performance level. 11.The wireless BS or UE of claim 9, wherein the processor is furtherconfigured to: use a blind CPE compensation method for subsequentlyreceived FDM symbols that are without embedded pilot symbols as long asthe control messages continue to indicate an adequate CPE compensationperformance level.
 12. The wireless BS or UE of claim 9, wherein theprocessor is further configured to: compensate each of the equalized FDMdata subcarriers using the CPE estimate.
 13. The wireless BS or UE ofclaim 9, wherein the other wireless BS or UE is enabled by the controlmessages to adapt the density of the embedded pilot symbols in a dynamicmanner.
 14. The wireless BS or UE of claim 9, wherein the other wirelessBS or UE is enabled by the control messages to adapt the density of theembedded pilot symbols in a semi-static manner.
 15. The wireless BS orUE of claim 9, wherein the processor generates the CPE estimate based ona constellation diagram associated with a modulation scheme used by theother wireless BS or UE to generate the FDM data subcarriers of thetransmitted FDM symbol.
 16. The wireless BS or UE of claim 9, whereinthe control messages that indicate the CPE compensation performancelevel comprise RRC control messages.
 17. A non-transitorycomputer-readable medium having instructions stored thereon that arecapable of causing or configuring a wireless base station or userequipment (BS or UE) to perform operations comprising: receiving, by thewireless BS or UE, a frequency division multiplexed (FDM) symboltransmitted by another wireless BS or UE; processing, by the BS or UE,the received FDM symbol to obtain its equalized FDM data subcarriers;generating, by the BS or UE, a CPE estimate using the equalized FDM datasubcarriers; sending, by the wireless BS or UE to the other wireless BSor UE, control messages that indicate a CPE compensation performancelevel using the CPE estimate; and wherein the other wireless BS or UE,in response to the control messages, is enabled to adapt a density intime and/or frequency of embedded pilot symbols within FDM symbolssubsequently transmitted to the wireless BS or UE.
 18. Thenon-transitory computer-readable medium of claim 17, the operationsfurther comprising: using, by the wireless BS or UE, a blind CPEcompensation method for subsequently received FDM symbols that arewithout embedded pilot symbols as long as the control messages continueto indicate an adequate CPE compensation performance level.
 19. Thenon-transitory computer-readable medium of claim 17, the operationsfurther comprising: compensating, by the BS or UE, each of the equalizedFDM data subcarriers using the CPE estimate.
 20. The non-transitorycomputer-readable medium of claim 17, wherein the control messages thatindicate the CPE compensation performance level comprise RRC controlmessages.